This section provides background information related to the present disclosure which is not necessarily prior art.
Equipment operated at very high output power levels (e.g. >3 kilowatts (KW)) typically uses high voltage input feeds, for maintaining current drawn from an AC utility power-line at practical levels. The utility power-lines typically have a 230 volt (V) 3-phase input or 380V, 400V, 415V, or 480V 3-phase input, depending on the application and the geographic location. Certain applications may also operate on a single-phase but with a preferred high input voltage of 200V or more.
Known high voltage power converters are typically designed for a particular input voltage and do not accommodate a wide range of input voltages. This results in multiple product designs and manufacturing versions to accommodate the many different input voltages required around the world. The need for these multiple product configurations increases development costs, increases inventory parts requirements, and complicates the calculus for determining the inventory requirements for finished products.
Further, most 3-phase power factor correction topologies were originally designed and developed for motor control applications that inherently have undesirable high frequency swinging voltage nodes with respect to earth-ground. The swinging nodes cause unacceptably high common mode noise making operation at the higher frequencies demanded by many current applications difficult. For example, most 3-phase power factor correction (PFC) topologies, including a Vienna Rectifier, have a high voltage bus that swings, with respect to earth, at a rate of the switching frequency. This voltage swinging is a major source of common mode emissions where the common mode currents get coupled to earth through a direct-current to direct-current (DC-DC) converter transformer's parasitic capacitance. This makes it very difficult to manage electromagnetic interference (EMI); ultimately requiring a reduced switching frequency and/or use of a bulky EMI filter.
Some designers prefer to use three independent single phase AC-DC converters connected in parallel at the output with a current sharing mechanism. Such designs use traditional single phase PFC topologies which deliver excellent harmonic rejections, high PFC, and a stable DC link bus. This approach reduces design complexity by using simple, proven design blocks. However, each AC-DC converter may need to operate with a 480V±10% AC input feed requiring a PFC boost converter to deliver an 800V DC link. Using known boost PFC approaches requires a boost switch and diode rating of more than 1000V, which are expensive compared to more common lower voltage switches and diodes. If the expensive 1000V devices are not used, either the converter performance will be compromised or a more complex converter design using high-performance 600V devices is needed. In addition, if the converters are needed to accommodate a 230V, Delta 3-phase input and a 380V-480V, Delta 3-phase input, the design is further complicated by the wide voltage range needs. It is well known that boosting over a large range deteriorates a converter's efficiency and significantly increases the manufacturing cost of the power converter. Such wide voltage range power supplies are employed for some products that can accept the increased cost and efficiency penalty. For example, user of mobile electronic devices, such as cell phones or notebook computers may travel across the globe requiring the devices' chargers and power sources to be compatible with worldwide utility feeds.
One known 3-phase delta input power supply uses three independent single phase, isolated power supplies, each having its own PFC and DC-DC converter block connected in parallel at output and deploying active or passive current sharing. One known single phase rail, shown at FIG. 1, uses inexpensive, high performance 600V devices and generates a boosted 800V DC link bus.
As seen in FIG. 1, a PFC converter 100 has a split DC link generating, for example 800V, with each capacitor, 102, 104 sharing 400V each. The boost switches 106, 108 are typically driven by identical signals through an appropriate isolation drive circuit. If the inductors 110, 112 do not have an exactly equal inductance value, the center node 114 voltage will vary from a center point of the input signal but still be at a fixed reference. However, this is possible only in ideal conditions when the two switches 106, 108 are turning on and off at the same time and at the same speed. The rise and fall of voltage across the two switches must be identical. In practice, the drive signals for the two switches are different due to variations in the switches' gate threshold, drive signal imbalance, layout parasitics, etc. Thus, because the two switches do not turn on and off at the same time, in a synchronized manner, the center node 114 voltage swings at every high frequency switching cycle. The extent of swing depends on the extent of variation in delays and the difference in rise and fall times of the two switches. The common mode EMI performance typically varies from unit-to-unit and is affected by temperature variations impacting the gate threshold of the two switches. Also this configuration does not allow use of a common core inductor for 110, 112. Further, if the variation in delays and switching times are large, the voltage on the two series bulk capacitors 102,104 will not be identical, requiring a special control scheme. Power converter 100 also includes an EMI filter shown generally at 116 and a bridge rectifier shown generally at 118.
The design of FIG. 1 is marginally improved using a pair of input splitting capacitors to reduce the swinging center node of the bulk capacitors, as shown in FIG. 2.
The control methodology for power converter 200 of FIG. 2 is essentially the same as described above, with respect to FIG. 1. If there is a short delay between the switching of 106, 108, the splitting capacitors 202, 204 maintain center node 114 at a fixed DC or low frequency level. Assuming that 106 turns off slightly before 108, inductor 110 starts freewheeling through diode 206, capacitor 102, switch 108, and inductor 112. Since 108 is still on, the current in inductor 112 continues to ramp by drawing current from capacitor 204 until 108 turns off. Due to a charge on capacitor 208, there is minimal swing at the center node 114. Similar action occurs when 106 turns on before 108. Current from 202 is drawn to ramp the current in 110. However, if there is a variation in switching times, the effective high frequency stability of center node 114 depends on the stiffness of the voltage across 202 and 204. The value of capacitors 202, 204 is dictated by the level of PFC needed. Very high values of 202, 204 deteriorate the displacement factor; while low values are subject to common mode EMI issues. Also, the power converter 200 does not allow use of a common core inductor for 110, 112 because of high frequency ringing caused by the leakage inductance of the coupled choke and capacitors 202, 204.